In order to remove the DC component of a signal or to eliminate DC offset, most integrated circuits rely on high-pass filters. Most commonly used examples of a high-pass filter are DC level shifters, DC blocking capacitors, and DC servo-loops.
Recently, there is much interest to develop instrumentation amplifiers with low-power dissipation, low-noise, high common mode rejection, and large input impedance for example towards the field of ambulatory biomedical signal monitoring. The large DC component of the biomedical signals necessitates the implementation of high-pass filter characteristics to the instrumentation amplifier.
These requirements lead to the usage of chopper-stabilized instrumentation amplifiers with high-pass filter characteristics. The most power efficient way of implementing high-pass filter characteristics to the chopper stabilized amplifiers is to use DC servo-loops [1], [2]. The DC servo-loop senses the DC level of the output and subtracts it from the input of the amplifier.
The previous implementations of the chopper stabilized amplifiers incorporating a DC servo-loop for implementing high-pass filter characteristics either subtract voltage, [2], or current, [1], from the input signal before it is amplified by the instrumentation amplifier. However, the prior technique results in large power dissipation due to the fact that the DC servo-loop must be capable of supplying wide range of current output, and the later reduces the input impedance of the amplifier.
In an attempt to reduce the power dissipation of a DC servo-loop that subtracts current from the input signal, a coarse servo-loop and a fine servo-loop can be used. The coarse servo-loop has discrete output levels where as the fine servo-loop has a continuous output range. Therefore, the output range of the fine servo-loop in the prior implementation can be reduced while the coarse-fine servo-loop is still capable of supplying the same output range.
However, the implementation of the coarse amplifier is critical such that the time constant of the output of the coarse servo-loop must be much slower than the (1/2πfHP), where fHP is the high-pass filter cut-off frequency of the instrumentation amplifier that is defined by the fine servo-loop. In this case, the effect of the changing coarse servo-loop to the output of the instrumentation amplifier can be minimized by the fine servo-loop. Therefore, the crucial need is to implement a large-time constant inside the coarse servo-loop so that the output of the coarse servo-loop changes very slowly, when it updates its output.
Large time constant implementations are presented in FIG. 1A, FIG. 1B and FIG. 1C. FIG. 1A uses an on-chip resistor, R1A, connected between the input, Vin1A, and the output, Vout1A, and an on-chip capacitor, C1A, connected between the output and the ground. However, considering the necessity for a high-pass filter cut-off frequency in the range of 0.1 Hz to 1 Hz for most of the biomedical signal acquisition systems, the time constant of this architecture must be set much lower than 1/(2π(0.1 Hz)), which leads to extremely large silicon area if implemented using architecture of FIG. 1A.
FIG. 1B shows the realization of a large time constant where the resistors are implemented by using pseudo resistors [3]. FIG. 1C shows the realization of a large time constant where the resistors are implemented using the structure proposed in [4]. Both approaches can occupy very small area, however, they suffer from the fact that the input voltage swing is only limited to less than hundred millivolts. This is due to the fact that the PMOS transistors of FIG. 1B and FIG. 1C operate in weak inversion, and a large input signal will increase the source-to-gate voltage of the PMOS transistors. This further turns on the transistor, leading to reduced time constant. Therefore, both the implementation of FIG. 1B and FIG. 1C needs the series connection of large number of transistors in order to improve their input range. This can lead to large silicon area consumption.